I. Field of the Invention
This invention relates generally to the field of wireless communications, and, more specifically, to a multi-band transceiver for a wireless communication device or handset.
II. Background
Wireless communication systems are an integral component of the ongoing technology revolution. Mobile radio communication systems, such as cellular telephone systems, are evolving at an exponential rate. In a cellular system, a coverage area is divided into a plurality of xe2x80x9ccellsxe2x80x9d. A cell is the coverage area of a base station or transmitter. Low power transmitters are utilized, so that frequencies used in one cell can also be used in cells that are sufficiently distant to avoid interference. Hence, a cellular telephone user, whether mired in traffic gridlock or attending a meeting, can transmit and receive phone calls so long as the user is within a xe2x80x9ccellxe2x80x9d served by a base station.
Mobile cellular systems were originally developed as analog systems. After their introduction for commercial use in the early 1980s, mobile cellular systems began to experience rapid and uncoordinated growth. In Europe, for example, individual countries developed their own systems. Generally, the systems of individual countries were incompatible, which constricted mobile communications within national boundaries and restricted the market for mobile equipment developed for a particular country""s system. In 1982, in order to address this growing problem, the Conference of European Posts and Telecommunications (CEPT) formed the Groupe Speciale Mobile (GSM) to study and develop a set of common standards for a future pan-European cellular network. It was recommended that two blocks of frequencies in the 900 MHz range be set aside for the system. The initial goals for the new system included international roaming ability, good subjective voice quality, compatibility with other systems such as the Integrated Services Digital Network (ISDN), spectral efficiency, low handset and base station costs, and the ability to support new services and a high volume of users.
One of the initial, major decisions in the development of the GSM standard was adoption of a digital, rather than an analog, system. As mentioned above, analog systems were experiencing rapid growth and the increasing demand was straining the capacity of the available frequency bands. Digital systems offer improved spectral efficiency and are more cost efficient. The quality of digital transmission is also superior to that of analog transmission. Background sounds such as hissing and static and degrading effects such as fadeout and cross talk are largely eliminated in digital systems. Security features such as encryption are more easily implemented in a digital system. Compatibility with the ISDN is more easily achieved with a digital system. Finally, a digital approach permits the use of Very Large Scale Integration (VLSI), thereby facilitating the development of cheaper and smaller mobile handsets.
In 1989, the European Telecommunications Standards Institute (ETSI) took over responsibility for the GSM standards. In 1990, phase I of the standard was published and the first commercial services employing the GSM standard were launched in 1991. It was also renamed in 1991 as the Global System for Mobile Communications (still GSM). After its early introduction in Europe, the standard was elevated to a global stage in 1992 when introduced in Australia. Since then, GSM has become the most widely adopted and fastest growing digital cellular standard, and is positioned to become the world""s dominant cellular standard. With (currently) 324 GSM networks in operation in 129 countries, GSM provides almost complete global coverage. As of January 1999, according to the GSM Memorandum of Understanding Association, GSM accounted for more than 120 million subscribers. Market research firms estimate that by 2001, there will be more than 250 million GSM subscribers worldwide. At that time, GSM will account for almost 60% of the global cellular subscriber base, with yearly shipments exceeding 100 million phones.
Two frequency bands of 25 MHz were allocated for GSM use. As illustrated in FIG. 1a, the 890-915 MHz band is reserved for transmission or xe2x80x9cuplinkxe2x80x9d (mobile station to base station), and the 935-960 MHz band is reserved for reception or xe2x80x9cdownlinkxe2x80x9d (base station to mobile station). An extra ten MHz of bandwidth was later added to each frequency band. The standard incorporating this extra bandwidth (two 35 MHz bands) is known as Extended GSM (EGSM). In EGSM, the transmission band covers 880-915 MHz and the receiving band covers 925-960 MHz (FIG. 1b). The terms GSM and EGSM are used interchangeably, with GSM sometimes used in reference to the extended bandwidth portions (880-890 MHz and 925-935 MHz). Sometimes, the originally specified 890-915 MHz and 935-960 MHz bands are designated Primary GSM (PGSM). In the following description, GSM will be used in reference to the extended bandwidth (35 MHz) standard.
Due to the expected widespread use of GSM, capacity problems in the 900 MHz frequency bands were anticipated and addressed. ETSI had already defined an 1800 MHz variant (DCS or GSM 1800) in the first release of the GSM standard in 1989. In DCS, the transmission band covers 1710-1785 MHz and the receiving band covers 1805-1880 MHz (FIG. 1c). In the United States, the Federal Communications Commission (FCC) auctioned large blocks of spectrum in the 1900 MHz band, aiming to introduce digital wireless networks to the country in the form of a mass market Personal Communication Service (PCS). The GSM service in the US is known as PCS or GSM 1900. In PCS, the transmission band covers 1850-1910 MHz and the receiving band covers 1930-1990 MHz (FIG. 1d).
Regardless of which GSM standard is used, once a mobile station is assigned a channel, a fixed frequency relation is maintained between the transmit and receive frequency bands. In GSM (900 MHz), this fixed frequency relation is 45 MHz. If, for example, a mobile station is assigned a transmit channel at 895.2 MHz, its receive channel will always be at 940.2 MHz. This also holds true for DCS and PCS; the frequency relation is just different. In DCS, the receive channel is always 95 MHz higher than the transmit channel and, in PCS, the receive channel is 80 MHz higher than the transmit channel. This frequency differential will be referred to in the ensuing discussion as the frequency offset.
The architecture of one implementation of a GSM network 20 is depicted in block form in FIG. 2. GSM network 20 is divided into four interconnected components or subsystems: a Mobile Station (MS) 30, a Base Station Subsystem (BSS) 40, a Network Switching Subsystem (NSS) 50 and an Operation Support Subsystem (OSS) 60. Generally, MS 30 is the mobile equipment or phone carried by the user; BSS 40 interfaces with multiple MSs 30 and manages the radio transmission paths between the MSs and NSS 50; NSS 50 manages system switching functions and facilitates communications with other networks such as the PSTN and the ISDN; and OSS 60 facilitates operation and maintenance of the GSM network.
Mobile Station 30 comprises Mobile Equipment (ME) 32 and Subscriber Identity Module (SIM) 34. ME 32 is typically a digital mobile phone or handset. SIM 34 is a memory device that stores subscriber and handset identification information. It is implemented as a smart card or as a plug-in module and activates service from any GSM phone. Among the information stored on SIM 34 are a unique International Mobile Subscriber Identity (IMSI) that identifies the subscriber to system 20, and an International Mobile Equipment Identity (IMEI) that uniquely identifies the mobile equipment. A user can access the GSM network via any GSM handset or terminal through use of the SIM. Other information, such as a personal identification number (PIN) and billing information, may be stored on SIM 34.
MS 30 communicates with BSS 40 across a standardized xe2x80x9cUmxe2x80x9d or radio air interface 36. BSS 40 comprises multiple base transceiver stations (BTS) 42 and base station controllers (BSC) 44. A BTS is usually in the center of a cell and consists of one or more radio transceivers with an antenna. It establishes radio links and handles radio communications over the Um interface with mobile stations within the cell. The transmitting power of the BTS defines the size of the cell. Each BSC 44 manages multiple, as many as hundreds of, BTSs 42. BTS-BSC communication is over a standardized xe2x80x9cAbisxe2x80x9d interface 46, which is specified by GSM to be standardized for all manufacturers. The BSC allocates and manages radio channels and controls handovers of calls between its BTSs.
The BSCs of BSS 40 communicate with network subsystem 50 over a GSM standardized xe2x80x9cAxe2x80x9d interface 51. The A interface uses an SS7 protocol and allows use of base stations and switching equipment made by different manufacturers. Mobile Switching Center (MSC) 52 is the primary component of NSS 50. MSC 52 manages communications between mobile subscribers and between mobile subscribers and public networks 70. Examples of public networks 70 that MSC 52 may interface with include Integrated Services Digital Network (ISDN) 72, Public Switched Telephone Network (PSTN) 74, Public Land Mobile Network (PLMN) 76 and Packet Switched Public Data Network (PSPDN) 78.
MSC 52 interfaces with four databases to manage communication and switching functions. Home Location Register (HLR) 54 contains details on each subscriber residing within the area served by the MSC, including subscriber identities, services to which they have access, and their current location within the network. Visitor Location Register (VLR) 56 temporarily stores data about roaming subscribers within a coverage area of a particular MSC. Equipment Identity Register (EIR) 58 contains a list of mobile equipment, each of which is identified by an IMEI, which is valid and authorized to use the network. Equipment that has been reported as lost or stolen is stored on a separate list of invalid equipment that allows identification of subscribers attempting to use such equipment. The Authorization Center (AuC) 59 stores authentication and encyrption data and parameters that verify a subscriber""s identity.
OSS 60 contains one or several Operation Maintenance Centers (OMC) that monitor and maintain the performance of all components of the GSM network. OSS 60 maintains all hardware and network operations, manages charging and billing operations and manages all mobile equipment within the system.
The GSM transmitting and receiving bands are divided into 200 kHz carrier frequency bands. Using Time Division Multiple Access techniques (TDMA), each of the carrier frequencies is subdivided in time into eight time slots. Each time slot has a duration of approximately 0.577 ms, and eight time slots form a TDMA xe2x80x9cframexe2x80x9d, having a duration of 4.615 ms. One implementation of a conventional TDMA frame 80 having eight time slots 0-7 is illustrated in FIG. 3.
In this conventional TDMA framework, each mobile station is assigned one time slot for receiving data and one time slot for transmitting data. In TDMA frame 80, for example, time slot zero has been assigned to receive data and time slot four has been assigned to transmit data. The receive slot is also referred to as the downlink slot and the transmit slot is referred to as the uplink slot. After the eight slots, the remaining slots are used for offset, control, monitoring and other operations. This framework permits concurrent reception by as many as eight mobile stations on one frequency and concurrent transmission by as many as eight mobile stations on one frequency.
As described above, there are currently three GSM frequency bands defined. With the proliferation of wireless handset usage showing now signs of slowing down, it is likely that additional bands will be defined in the future. Hence, GSM mobile stations intended for global usage should have multi-band capability. Unfortunately, because of the widely disparate frequency ranges of the GSM, DCS, and PCS systems, a transceiver with a single main oscillator has not been able to cover the required frequencies. Moreover, designs employing separate oscillators for each of the bands are not feasible because of the cost involved, while designs employing a single switchable oscillator typically suffer from poor performance.
Another problem is that current multi-band handsets utilize off-chip components such as the receiver""s intermediate frequency (IF) filter which, in one conventional design, comprises a surface acoustical wave (SAW) filter. Components such as this tend to be large and bulky, and consume excessive space. Thus, they are inconsistent with subscribers"" demand for handsets are as compact, lightweight, and mobile as possible.
Direct conversion receivers eliminate the need for IF filters. However, current direct conversion receivers are susceptible to self-conversion to DC of the local oscillator signal or large RF blockers.
This problem can be further explained with reference to FIG. 4, which illustrates a conventional direct conversion receiver. As illustrated, the receiver of FIG. 4 comprises an antenna 200 coupled to the radio frequency (xe2x80x9cRFxe2x80x9d) input port 219 of mixer 211. Mixer 211 has a local oscillator (xe2x80x9cLOxe2x80x9d) input port 214, and an output port 201. The mixer mixes the signals provided at the RF and LO input ports, and provides the mixed signal to the output port. In the receiver of FIG. 4, the frequency of the signal provided at the LO input port, fLO, is matched to the frequency of the carrier frequency, fRF, of the signal provided at the RF input port such that fLO=fRF. The mixed signal provided at the output port 201 of mixer 211 has a first order component at the baseband frequency, fBB, and a first component at twice the LO or RF carrier frequencies, or 2fLO.
The output port 201 of mixer 214 is coupled to LPF 212 through signal line 213. The purpose of LPF 212 is to select only the baseband component of the signal output from mixer 211 while suppressing the higher frequency component at the frequency 2fLO. LPF 212 also rejects any unwanted signals outside the desired band around fBB. The output of the LPF 212 is provided on signal line 215. It represents the baseband portion of the RF signal received over antenna 200.
An advantage of the design of FIG. 4 is the elimination of the IF filter, and related components such as a second mixer. However, a problem with this design is its vulnerability to leakage between the signals on the RF and IF input ports of the mixer. This problem is explained further in the following section.
With reference to FIG. 4, consider the case in which a portion of the signal provided at the LO input port leaks onto the RF input port. Such is identified with reference numeral 216 in FIG. 4. This portion will be mixed by mixer 211 with the original LO signal, thus producing a distortion in the output signal at the baseband frequency. Since this distortion is at the baseband frequency, it will pass through LPF 212, and appear in the output signal provided on signal line 215. The result is that this output signal is distorted in relation to the original transmitted baseband signal.
Consider next the case in which a portion of the signal provided at the RF input port leaks onto the LO input port. Such is represented by identifying numeral 217 in FIG. 4. This portion will be mixed by mixer 211 with the original RF signal, thus producing a distortion in the output of the mixer at the baseband frequency. Again, this distortion, being at the baseband frequency, will appear in the output signal provided on signal line 215.
In addition to leakage between the RF and LO input ports, another problem stems from the LO signal leaking onto and being radiated by antenna 200. This leakage is represented by identifying numeral 218 in FIG. 4. This leakage can interfere with other similar receivers that may be present in the same geographical area since the radiated LO component is at the same frequency as the RF signals received by these other receivers.
This leakage problem renders the direct conversion receiver of FIG. 4 unsuitable for use in applications such as GSM mobile wireless handsets, and other systems with large blocker suppression requirements, because the distortion introduced by the leakage is unacceptable for these applications.
Efforts to solve this problem have involved shielding and physical separation between the RF and LO inputs. Shielding, however, is expensive and often ineffective at the high frequencies which typically characterize current mobile wireless phones, 900 MHz or more. Moreover, physical separation is impractical for use in wireless handsets, in which space is at a premium. Port to port isolation of the mixer is also a finite value which typically becomes less at higher frequencies.
The distortion introduced by leakage always results in unwanted DC at the mixer output. For GSM and some other systems, this DC is not allowed to be removed by mechanisms such as a blocking capacitor because the desired signal may itself contain DC.
Accordingly, there is a need for a multi-band transceiver which overcomes the disadvantages of the prior art.
In accordance with the purpose of the invention as broadly described herein, there is provided a multi-band transceiver for transmitting and receiving RF signals in one of a plurality of frequency bands. Advantageously, the transceiver is configured for use in a wireless communication device, whether a mobile device or handset, or a base station or other infrastructure component. In one implementation, the transceiver is configured for the GSM and DCS bands; in another, the GSM, DCS, and PCS bands.
The receiver portion of the transceiver includes a direct conversion receiver (DCR). A signal derived from a tunable local oscillator services the receiver. In addition, in one embodiment, the local oscillator is shared with an upconverter in the transmitter portion of the transceiver.
The direct conversion receiver includes a frequency translator having first and second ports. In one implementation, the frequency translator is a mixer. In another, it is a multiplier. A first filter is coupled to the first port, and a second filter is coupled to the second port. Preferably, the filters are integral with or inherent to the ports so that the frequency translator lacks exposed unfiltered ports. A third filter is coupled to the output of the frequency translator. It is advantageously a low pass filter configured to provide as an output signal the baseband component of the signal output from the frequency translator.
In operation, one of the plurality of bands is selected. A signal derived from the output of the local oscillator is coupled to the first filtered port of the frequency translator of the DCR. The frequency f1 of the signal is set through suitable tuning of the local oscillator such that it is about an nth order subharmonic of the carrier frequency f2 of the signal that is to be applied to the second filtered port of the frequency translator of the DCR, wherein n is an integer greater than 1. That is to say, f1≅(1/n)f2, wherein n is an integer greater than 1. (For purposes of this disclosure, use of the terms such as xe2x80x9caboutxe2x80x9d or xe2x80x9capproximatelyxe2x80x9d or xe2x80x9csubstantiallyxe2x80x9d or the symbol xe2x80x9c≅xe2x80x9d for describing frequency or timing relationships between signals and the like is intended to take account of tolerances which are acceptable in the trade, and to allow some leeway in the description of these relationships which is consistent with these tolerances when strict mathematical exactitude may not be possible.)
The first filter is preferably a low pass filter having a corner frequency below the selected band, and above the frequency of the nth order harmonic. In other words, the corner frequency is above f1 and below f2. Consequently, it is configured to substantially attenuate the frequency f2 to the first unfiltered port of the frequency translator. Similarly, the second filter is preferably a high pass filter having a corner frequency below the selected band, and above the frequency of the nth order harmonic. Again, the corner frequency is above f1 and below f2. Consequently, it is configured to attenuate the first frequency f1 to the second unfiltered port of the frequency translator.
Through operation of these filters, the effects of leakage between the first and second ports of the frequency translator are eliminated or reduced. Leakage from the first port to the second port will be at the frequency f1, and thus attenuated by the second filter. Similarly, leakage from the second port to the first port will be at the frequency f2, and thus attenuated by the first filter. Thirdly, radiation at the frequency f1 out through the antenna will be blocked by a bandpass filter located upstream of the DCR which has a passband centered on the selected band.
In one embodiment, the frequency translator is a multiplier configured to multiply the signals at the first and second input ports thereof. In another embodiment, the frequency translator is a mixer configured to switch the second input to the output through a switching action which is performed at a switching or sampling rate of n times the frequency f1 of the signal applied to the first input of the mixer. By switching at n times the frequency f1, the mixer conserves frequency in that more energy is packed into the baseband component of the output of the mixer output than if the switching action were performed at the frequency f1.
In one embodiment, the transmitter portion of the transceiver comprises a modulator coupled to an upconverter. A carrier input source provides the carrier input to the modulator. The carrier input source comprises a frequency adjuster coupled to the output of the crystal oscillator providing the reference frequency to the phase locked loop which comprises the local oscillator. The frequency adjuster is configured to receive the output of the crystal oscillator, and to provide an output signal having a frequency which is equal to the frequency of the output of the crystal oscillator adjusted by a variable amount responsive to the selected frequency band. In one implementation, the frequency adjuster is a frequency multiplier.
In a second embodiment, the carrier input source comprises a frequency adjuster coupled to the output of the phase locked loop which comprises the local oscillator. The frequency adjuster is configured to receive the output of the phase locked loop, and to provide an output signal having a frequency which is equal to the frequency of the output of the phase locked loop adjusted by a variable amount responsive to the selected frequency band. In one implementation, the frequency adjuster is a frequency divider.
In a third embodiment, the modulator is within a loop of a translation loop upconverter, and the carrier input of the modulator is derived from a downconversion frequency translator included within the loop.
In one configuration, the modulator is a quadrature modulator, the upconverter is a translation loop upconverter, and the carrier input source is a low frequency offset source. In one implementation, the quadrature modulator and low frequency offset source are outside the loop of the translation loop upconverter. In a second implementation, the quadrature modulator and low frequency offset source are within the loop of the translation loop upconverter.
In the case of the first configuration, a low frequency offset source provides the carrier input to the quadrature modulator. The frequency of the carrier signal is a variable depending on the selected band. It is selected to be about equal to the offset frequency for the selected band, that is, the offset between the transmit and receive channels for the selected band. The translation loop upconverter includes a transmit downconversion frequency translator. The frequency translator is of the type which switches or samples at n times the frequency of the signal provided at the first input thereof. The value of n for this frequency translator is the same as that for the frequency translator in the DCR of the receiver portion of the transceiver.
In the case of the second configuration, the carrier signal for the quadrature modulator is derived from the output of the frequency translator in the translation loop. The loop is configured so that the frequency of the output of the frequency translator is, after suitable filtering, about equal to the frequency offset for the selected band. In that sense, the frequency translator functions as the low frequency offset source.
In both configurations, a low pass filter is inherent to or integral with the first input of the frequency translator, such that the unfiltered first input is covered and not exposed. The local oscillator in the receiver portion of the transceiver is shared with the frequency translator in the translation loop upconverter in that a signal derived from the local oscillator is coupled to the filtered first input of the frequency translator. In operation, the frequency of the signal applied to this input is about an nth order subharmonic of the signal applied to the second input of the frequency translator, wherein n is an integer greater than 1.
The translation loop upconverter in both configurations receives the output of the quadrature modulator and increases the carrier frequency of this output to about the appropriate frequency for transmission. This frequency is the frequency of the selected receive channel in the selected band minus the frequency offset for the selected band.
In one implementation, each of the foregoing frequency translators is a mixer, with the first input port being an LO input port, and the second input port being an RF input port. In this implementation, a procedure known as half-frequency injection is utilized. According to this procedure, the frequency f1 of the signal applied to the LO input port is xc2xd of the frequency f2, the carrier frequency of the RF signal applied to the second port.
In one implementation example, the transceiver is configured to handle the GSM and DCS bands. In this implementation, two switchable and selectable DCRs are provided. In operation, the DCR corresponding to the selected band is selected and switched such that it is in the signal path from the baseband filter to the switch/band selector. The first DCR is preceded by a bandpass filter having a passband defined by the GSM receive band, 925-960 MHz. The second DCR is preceded by a passband defined by the DCS receive band, 1805-1880 MHz. The local oscillator in this implementation is the output of a phase locked loop (PLL). The PLL includes a fractional N synthesizer. A reference divider at the output of a crystal oscillator at 13 MHz provides the reference frequency to the PLL. The output of the PLL is tunable in the range of 450.25 MHz to 480 MHz. The output of the PLL is applied to the LO input of the mixer in the first DCR. The output of the PLL is also passed through a doubler, and the output of the doubler is applied to the LO input of the mixer of the second DCR.
In the case in which the GSM band is selected, the PLL is tuned such that the frequency of the output thereof is about xc2xd the frequency of the selected channel in the GSM band. In the case in which the DCS band is selected, the PLL is tuned so that the output thereof is about xc2xc the frequency of the selected channel on the DCS band. This way, through action of the doubler, the signal which is applied to the LO input of the mixer in the DCR corresponding to the DCS band is about xc2xd the frequency of the selected channel in the DCS band.
In one configuration of this implementation, a transmitter portion of the transceiver includes a quadrature modulator followed by a translation loop upconverter. A low frequency offset source provides a carrier input to the quadrature modulator at a frequency about equal to the frequency offset between the receive and transmit channels for the selected band. As discussed, the frequency offset for the GSM band is 45 MHz, that for the DC band is 95 MHz, and that for the PCS band is 80 MHz.
In one example of this configuration, the carrier input is derived by multiplication of the crystal oscillator reference frequency by a multiplication factor which depends on the selected band. For the GSM band, assuming a 13 MHz crystal oscillator reference frequency, the multiplication factor is advantageously 3, yielding a carrier offset of 39 MHz. For the DCS band, again assuming a 13 MHz crystal oscillator reference frequency, the multiplication factor is advantageously 7, yielding a carrier offset of 91 MHz.
In another example of this configuration, the carrier input is derived by dividing the output of the PLL by a division factor which depends on the selected band. For the GSM band, assuming a PLL output frequency of 450-480 MHz, the division factor is advantageously 10, yielding a carrier offset in the range of 45-48 MHz. For the DCS band, again assuming a PLL output frequency of 450-480 MHz, the division factor is advantageously 5, yielding a carrier offset in the range of 90-96 MHz.
In a second configuration of this implementation, the quadrature modulator is contained within the loop of the translation loop upconverter in that the output of the downconversion mixer in the loop provides, after suitable filtering, the carrier input of the quadrature modulator. The loop is configured such that the carrier input to the quadrature modulator is about the frequency offset for the selected band.
In both configurations, the translation loop upconverter is configured to increase the carrier frequency of the output of the quadrature modulator so that it is at the appropriate frequency for transmission. In the case of DCS, the transmit band is 1710-1785 MHz. In the case of GSM, the transmit band is 890-915 MHz. The appropriate frequency for transmission is the selected channel within the appropriate transmit band, which has a frequency equal to that of the selected channel in the receive band minus the frequency offset for the band.
In both configurations, the output of the PLL is shared by the translation-loop upconverter in that a signal derived from the output from the PLL is provided to the filtered LO input of the downconversion mixer in the translation loop upconverter. In the case of the GSM band, the PLL output is applied directly to the filtered LO input of the mixer. In the case of the DCS band, the PLL output, after passage through the doubler, is applied to the LO input of the mixer.
A related method of providing full duplex transmission and reception is provided which comprises the following steps: selecting a band from a plurality of bands; receiving a signal at a channel within the selected band, the channel having a frequency; directly converting the signal to a baseband signal using a first signal derived from a local oscillator signal, the first signal being an nth subharmonic of the channel frequency, wherein n is an integer greater than 1; upconverting a second baseband signal to a transmission frequency; and transmitting the upconverted signal.